In a digital communications system, the use of constant amplitude modulations, such as continuous phase modulations, is preferred because these allow the range of the transmitted signal to be maximized. This is because constant-envelope modulations have the benefit of allowing the transmitted signal to have an almost constant power. The phase continuity allows the signal to occupy a smaller bandwidth, and the constant envelope of the signal allows better resistance to the non-linearities of the transmission channel and allows the amplifiers of the system to operate close to their saturation point. A continuous-phase-modulated signal is likewise called a CPM signal.
The advantageous properties of continuous phase modulations appear to a greater extent when the modulation order used is high or when the length of the memory of the modulation is high. However, this results in great implementation complexity for the receiver.
In order to overcome this disadvantage, it is known practice to implement breakdown of the continuous-phase-modulated signal in the form of a sum of amplitude-modulated signals, each component of the form being defined from the parameters of the modulation and weighted by a pseudo-symbol determined from the symbols to be transmitted. The principle of such a breakdown is presented in article [1]. According to the teaching of this document, it is known practice to use the structure of the receiver described in FIG. 1 in order to demodulate a continuous-phase-modulated signal.
Such a receiver 100 has a filter bank 101, 102, . . . 10D for filtering the received signal SR. The filters of the bank are each defined by a temporal response C0(−t) . . . CD−1(−t) that is suited to the breakdown in the form of amplitude-modulated components according to [1]. The receiver 100 likewise has samplers 111, 112, . . . 11D for sampling the filtered signals and a demodulator 120 that executes an algorithm based on a trellis, for example a Viterbi algorithm or a BCJR (Bahl, Cocke, Jelinek and Raviv) algorithm or any other algorithm of the same type. The demodulator 120 determines the most likely transmitted symbol from the filtered symbols. The states and the metrics used by the trellis are developed in document [2].
The receiver 100 described in FIG. 1 has several disadvantages, however.
Firstly, the components of the breakdown of the signal as a sum of amplitude-modulated signals are not orthogonal with respect to one another. Thus, the filters 101, 102, . . . 10D that are suited to these components are not orthogonal either, which gives rise to intercomponent interference on the demodulated signal SD.
Secondly, the aforementioned components have a duration that can extend beyond the duration of a symbol to be transmitted. Thus, over each symbol period, the contributions of a plurality of symbols can interfere, the filters of the receiver 100 therefore introducing intersymbol interference.
Thirdly, the samples of the noise at the output of the filters 101, 102, . . . 10D whose duration is greater than the duration of a symbol are correlated. The noise at the output of the filter bank therefore cannot be considered to be white Gaussian noise. Branch metrics of the trellis that are used in the Viterbi algorithm or in the BCJR algorithm are then no longer exact, since these are constructed on the strict assumption of additive white Gaussian noise.
The known solutions allowing the three aforementioned disadvantages to be resolved consist most often either in disregarding the effects thereof or in introducing signal equalization solutions and/or noise whitening filters into the receiver. The addition of these solutions considerably increases the complexity of implementation of the receiver.